The present invention relates to an amplifier circuit and more particularly, to a variable gain circuit, a wideband circuit and a gain control circuit suitable for a video amplifier circuit for a CRT display.
A conventional video amplifier circuit for a color CRT display is shown in FIG. 3. In the circuit shown in FIG. 3, video signals for the three primary colors of R(red), G(green) and B(blue) are applied from signal sources 1R, 1G and 1B through video amplifier circuits 2R, 2G and 2B to cathodes 3R, 3G and 3B of a CRT display 4, respectively. The CRT 4 includes a first grid 40 shared by the signals of the three primary colors RGB.
Since the conventional amplifier circuits for the three primary colors have the same circuit configuration, description of the operation of only the amplifier circuit for the red color R is being presented for purposes of explanation. An input signal voltage V.sub.IR is applied to a base of a transistor 22R constituting a grounded-emitter amplifier circuit and is amplified by the transistor so that an inverted output signal voltage V.sub.OR is produced at its collector and is supplied to the cathode 3R of the CRT 4, An operation point of the output signal voltage V.sub.OR in this case is adjusted by using a variable resistor 27R for adjustment of cut-off, and a voltage gain is adjusted by using a variable resistor 23R for adjustment of drive. A range of adjustment for the operation point of the output signal voltage is limited by a resistor 26R. A resistor 24R limits the voltage gain adjustment range in the same manner as the resistor 26R and also compensates for the frequency characteristic of the video amplifier circuit 2R at the same time. The frequency band of the video amplifier circuit 2R is mainly limited by a load capacitance 30 and a collector resistor 21R and is 1/(2.pi.C.sub.LR R.sub.CR) when the frequency characteristic is not compensated for. The frequency characteristic is improved in excess of the frequency determined by a time constant defined by the resistor 24R and an emitter peaking capacitor 25R added to suppress the band limitation. However, the upper limit of the frequency band increased by the improved frequency characteristic is reduced in inverse proportion to increase of a signal amplitude.
Adjustment of the white balance in the prior art is attained by repeatedly performing the cut-off adjustment of the primary color circuits and the drive adjustment of the video amplifier circuit for at least two primary colors.
An example of a conventional wideband circuit is now shown in FIG. 14. The circuit shown in FIG. 14 is named a cascode amplifier circuit and is advantageous in that the Miller effect occurring in a grounded-emitter transistor 22 of a first stage is suppressed. More particularly, with a mere grounded-emitter configuration, a collector voltage of the transistor 22 is swung to a voltage gain Av times a base voltage in an inverted direction. Generally, a signal source 1 is regarded as a series circuit of a signal voltage source 1v and a signal source resistor 1r. Accordingly, an input capacitance of a base terminal of the transistor 22 is substantially equal to a value of (1+Av) times a value C.sub.BC2 of a capacitance 210 between the base and the collector thereof and a time constant on the side of the input terminal determined by the capacitance value of the base terminal and a value rs of the signal source resistor 1r is increased so that the frequency band of the circuit is narrowed. Thus, as shown in FIG. 14, the load of the grounded-emitter circuit is formed by a grounded-base circuit having a low input impedance and constituted of a transistor 208, so that a signal voltage appearing at the collector of the transistor 22 is suppressed. The input capacitance of the base terminal of the transistor 22 is a sum of the capacitance C.sub.BC2 and an equivalent capacitance of a circuit including resistors 23 and 24 and a capacitor 25 connected to the emitter of the transistor 22 reflected to the base side of the transistor 22 through the transistor 22 and is a small value to an extent that the frequency characteristic is not adversely affected. Accordingly, the cascode amplifier circuit shown in FIG. 14 is widely applied to the video amplifier circuit and various wideband circuits including an integrated circuit as a circuit capable of spreading the frequency band sufficiently. In FIG. 14, a terminal 212 is an output terminal.
A wideband variable gain amplifier circuit is shown in FIG. 25A. In FIG. 25A, a signal current 98 converted by a transistor 92 and a resistor 97 from a voltage is divided by a fixed ratio set in transistors 93 and 94 constituting a differential pair to vary a gain. Thus, a signal Vi of an input signal source 1 is amplified and is outputted at output terminal 95 as output signal Vo. The division ratio in this case is constant regardless of a magnitude i.sub.s of the signal current 98 but, however, can be varied by adjusting a voltage .DELTA.V.sub.B of a differential pair control voltage source 9. Further, since the transistors 93 and 94 are configured as a grounded-base circuit in series connection with a grounded-emitter circuit constituted of the transistor 92, the circuit of FIG. 25A is considered to constitute a cascode amplifier circuit and can attain the wideband characteristic.
Adjustment process of the white balance in the prior art is described with reference to an input and output characteristic diagram, of a video amplifier circuit, shown in FIG. 4. In the characteristic diagram of FIG. 4, the abscissa represents an input signal voltage V.sub.I and the ordinate represents an output signal voltage V.sub.o. A straight line 50 shown by a solid line represents an input and output characteristic in the case where the white balance to be targeted is ensured.
In the current adjustment of the white balance, the cut-off adjustment and the drive adjustment are made in the state where input signal voltages of FIG. 4 are V.sub.IC V.sub.ID. If it is assumed that the video amplifier circuit has the characteristic shown by a broken line 51 at an initial state thereof, the output voltage V.sub.o is mainly shifted as shown by an arrow 52 by performing a first cut-off adjustment, so that the characteristic is shifted as shown by broken line 53. Adjustment of the voltage gain is made as shown by an arrow 54 in a next drive adjustment, so that the characteristic shown by a broken line 55 is obtained and a first adjustment of the white balance is finished. However, as apparent from an arrow 56 of FIG. 4, there is a problem that the precedingly performed cut-off adjustment is additionally shifted by the drive adjustment. Such interference between the cut-off adjustment and the drive adjustment in the prior art is now analyzed quantitatively. The video amplifier circuit for the red color R shown in FIG. 3 is taken up as a circuit example. When an output voltage at the time of the cut-off adjustment is V.sub.OCR, and V.sub.0 =V.sub.OCR, V.sub.1 =V.sub.CC, V.sub.2 =V.sub.ICR, V.sub.3 =V.sub.BE22R, V.sub.4 =V.sub.ER, R.sub.1 =R.sub.CR and R.sub.2 =R.sub.ER, the output voltage can be expressed as follows: ##EQU1## where V.sub.BE22R : voltage between the base and the emitter of the transistor 22R; and
V.sub.ER : emitter equivalent voltage ##EQU2## R.sub.ER : emitter equivalent resistance EQU R.sub.ER .apprxeq.(R.sub.23 +R.sub.24)//{(R.sub.26 +(R.sub.1R //R.sub.2R)}(3)
where R.sub.1 //R.sub.2R represents a parallel resistance value (R.sub.1R.R.sub.2R /(R.sub.1R +R.sub.2R)) of resistances R.sub.1R and R.sub.2R, and a parallel resistance value is hereinafter expressed thereby similarly.
Further, when a voltage gain set by the drive adjustment is A.sub.VR, the voltage gain is represented by ##EQU3##
From the foregoing, it would be understood that the resistances R.sub.1R and R.sub.2R Set in the cut-Off adjustment and the resistance R.sub.23 set in the drive adjustment exist together in the equations of V.sub.ER and R.sub.ER and the respective effects are preserved without canceling each other. Accordingly, since there is interference between the cut-off adjustment and the drive adjustment in the prior art, there is a problem that the adjustment of the white balance is not completed unless these adjustments are made repeatedly.
An example of the frequency characteristic at the input terminal of the cascode amplifier circuit of FIG. 14 is shown in FIG. 15. In the characteristic diagram of FIG. 15, the abscissa represents a signal frequency fs and the ordinate represents an amplitude .vertline.V.sub.B .vertline. of a signal at the base electrode of the transistor 22 which is an input terminal. As shown by a characteristic curve 213 of FIG. 15, a trap occurs in a frequency fr lower than a cut-off frequency f.sub.CB determined by the resistance r.sub.s of the signal source and the capacitance C.sub.BC2 between the base and the collector of the transistor 22. In order to make clear the cause for the occurrence of the trap, the transistors 208 and 22 are expressed by a hybrid .pi.-shaped equivalent circuit shown in FIG. 16 for analysis. In FIG. 16, r.sub.bb represents an inter base resistance, r.sub.E an emitter bulk resistance, r.sub..pi. an operation resistance between the base and the emitter, C.sub..pi. a diffusion capacitance, C.sub..mu. a capacitance between the base and the collector, C.sub.o a capacitance between the collector and the emitter, and g.sub.m a mutual conductance.
(1) An input impedance Z.sub.E as viewed from the emitter electrode of the grounded-base transistor 208 exhibits the inductivity as expressed by the following equation (5), where .beta..sub.0 is a low-frequency current amplification factor. ##EQU4##
(2) An input impedance Z.sub.B of the grounded-emitter transistor 22 is proportional to a series impedance of a parallel capacitance C.sub..mu. and C.sub.O and a load impedance Z.sub.C connected to the collector electrode as shown in the following equation (6): ##EQU5##
(3) From the above equations (1) and (2), it is considered that the trap occurs by a series resonance of an impedance component of the input impedance Z.sub.E of the transistor 208 and a parallel combined capacitance of the capacitance 210 between the base and the collector and a capacitance 211 of the collector and the emitter of the transistor 22. It has been confirmed by an experiment that the trap occurs in the vicinity of a resonance frequency fr expressed by the equation (7 ). However, all parameters of the equivalent circuit of the transistor in the equation (7) are values concerning the transistor 208. ##EQU6##
Since a waveform distortion occurs in the vicinity of the frequency fr in which the trap occurs, there is a problem that there is a case where the cascode amplifier circuit must suppress the frequency band at the sacrifice of its feature.
The temperature characteristic of the gain of the variable gain amplifier circuit shown in FIG. 25A will now be considered. The gain is proportional to the division ratio of the signal current 98 flowing through the transistors 93 and 94 constituting the differential pair, that is, a ratio i.sub.e1 /i.sub.e2 of emitter currents 290 and 291 and can be expressed as shown by the following equation (8): EQU i.sub.e1 /i.sub.e2 .apprxeq.1/{1+exp(q.DELTA.V.sub.B /kTj)}(8)
where g is an amount of electron charges, k is the Bolzmann constant, and T.sub.j is a temperature of the junction of the transistors 93 and 94.
Accordingly, in order to obtain a fixed gain, it is necessary to vary a temperature coefficient .differential..DELTA.V.sub.B /.differential.T.sub.j for a voltage .DELTA.V.sub.B of a differential pair control voltage source 9 in accordance with .DELTA.V.sub.B as shown by the characteristic 293 of FIG. 25B. A variation amount of the junction temperature T.sub.j is equal to a variation amount of the ambient temperature of the transistors 93 and 94. Accordingly, if the voltage .DELTA.V.sub.B is constant regardless of the ambient temperature, the ratio i.sub.e1 /i.sub.e2 of the divided currents is drifted toward 1 with the increase of the ambient temperature.